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Ultralow Distortion, Ultralow Noise Op Amp AD797
FEATURES
Low noise 0.9 nV/Hz typical (1.2 nV/Hz maximum) input voltage noise at 1 kHz 50 nV p-p input voltage noise, 0.1 Hz to 10 Hz Low distortion -120 dB total harmonic distortion at 20 kHz Excellent ac characteristics 800 ns settling time to 16 bits (10 V step) 110 MHz gain bandwidth (G = 1000) 8 MHz bandwidth (G = 10) 280 kHz full power bandwidth at 20 V p-p 20 V/s slew rate Excellent dc precision 80 V maximum input offset voltage 1.0 V/C VOS drift Specified for 5 V and 15 V power supplies High output drive current of 50 mA
PIN CONFIGURATION
OFFSET NULL 1 -IN 2 +IN 3 -VS 4 TOP VIEW
AD797
DECOMPENSATION AND DISTORTION NEUTRALIZATION 7 +VS
8 5
OFFSET NULL
Figure 1. 8-Lead Plastic Dual In-Line Package [PDIP] and 8-Lead Standard Small Outline Package [SOIC]
GENERAL DESCRIPTION
The AD797 is a very low noise, low distortion operational amplifier ideal for use as a preamplifier. The low noise of 0.9 nV/Hz and low total harmonic distortion of -120 dB at audio bandwidths give the AD797 the wide dynamic range necessary for preamps in microphones and mixing consoles. Furthermore, the AD797's excellent slew rate of 20 V/s and 110 MHz gain bandwidth make it highly suitable for low frequency ultrasound applications. The AD797 is also useful in infrared (IR) and sonar imaging applications, where the widest dynamic range is necessary. The low distortion and 16-bit settling time of the AD797 make it ideal for buffering the inputs to - ADCs or the outputs of high resolution DACs, especially when the device is used in critical applications such as seismic detection or in spectrum analyzers. Key features such as a 50 mA output current drive and the specified power supply voltage range of 5 V to 15 V make the AD797 an excellent general-purpose amplifier.
APPLICATIONS
Professional audio preamplifiers IR, CCD, and sonar imaging systems Spectrum analyzers Ultrasound preamplifiers Seismic detectors - ADC/DAC buffers
5
-90
INPUT VOLTAGE NOISE (nV/Hz)
4
-100
00846-001
6
OUTPUT
0.001
THD (dB)
-110
0.0003
2
-120
0.0001 MEASUREMENT LIMIT
1
10
100
1k
10k
100k
1M
10M
FREQUENCY (Hz)
00846-002
300
1k
3k 10k FREQUENCY (Hz)
30k
100k
300k
Figure 2. AD797 Voltage Noise Spectral Density
Figure 3. THD vs. Frequency
Rev. F
Information furnished by Analog Devices is believed to be accurate and reliable. However, no responsibility is assumed by Analog Devices for its use, nor for any infringements of patents or other rights of third parties that may result from its use. Specifications subject to change without notice. No license is granted by implication or otherwise under any patent or patent rights of Analog Devices. Trademarks and registered trademarks are the property of their respective owners.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A. Tel: 781.329.4700 www.analog.com Fax: 781.461.3113 (c)2008 Analog Devices, Inc. All rights reserved.
00846-003
0
-130 100
THD (%)
3
AD797 TABLE OF CONTENTS
Features .............................................................................................. 1 Applications....................................................................................... 1 Pin Configuration............................................................................. 1 General Description ......................................................................... 1 Revision History ............................................................................... 2 Specifications..................................................................................... 3 Absolute Maximum Ratings............................................................ 5 ESD Caution.................................................................................. 5 Typical Performance Characteristics ............................................. 6 Theory of Operation ...................................................................... 11 Noise and Source Impedance Considerations........................ 11 Low Frequency Noise ................................................................ 12 Wideband Noise ......................................................................... 12 Bypassing Considerations ......................................................... 12 The Noninverting Configuration............................................. 13 The Inverting Configuration .................................................... 14 Driving Capacitive Loads.......................................................... 14 Settling Time............................................................................... 14 Distortion Reduction ................................................................. 15 Outline Dimensions ....................................................................... 18 Ordering Guide .......................................................................... 19
REVISION HISTORY
1/08--Rev. E to Rev. F Changes to Absolute Maximum Ratings ....................................... 5 Change to Equation 1..................................................................... 12 Changes to the Noninverting Configuration Section................ 13 Updated Outline Dimensions ....................................................... 19 Changes to Ordering Guide .......................................................... 20 7/05--Rev. D to Rev. E Updated Figure 1 Caption ............................................................... 1 Deleted Metallization Photo ........................................................... 6 Changes to Equation 1 ................................................................... 12 Updated Outline Dimensions ....................................................... 19 Changes to Ordering Guide .......................................................... 20 10/02--Rev. C to Rev. D Deleted 8-Lead CERDIP Package (Q-8)..........................Universal Edits to Specifications ...................................................................... 2 Edits to Absolute Maximum Ratings ............................................. 3 Edits to Ordering Guide .................................................................. 3 Edits to Table I .................................................................................. 9 Deleted Operational Amplifiers Graphic .................................... 15 Updated Outline Dimensions ....................................................... 15
Rev. F | Page 2 of 20
AD797 SPECIFICATIONS
TA = 25C and VS = 15 V dc, unless otherwise noted. Table 1.
Parameter INPUT OFFSET VOLTAGE Offset Voltage Drift INPUT BIAS CURRENT TMIN to TMAX INPUT OFFSET CURRENT OPEN-LOOP GAIN TMIN to TMAX VOUT = 10 V RLOAD = 2 k TMIN to TMAX RLOAD = 600 TMIN to TMAX @ 20 kHz 1 G = 1000 G = 1000 2 G = 10 VOUT = 20 V p-p, RLOAD = 1 k RLOAD = 1 k 10 V step VCM = CMVR TMIN to TMAX VS = 5 V to 18 V TMIN to TMAX f = 0.1 Hz to 10 Hz f = 10 Hz f = 1 kHz f = 10 Hz to 1 MHz f = 1 kHz 5 V, 15 V 15 V 1 1 1 1 14,000 Conditions TMIN to TMAX 5 V, 15 V 5 V, 15 V Supply Voltage (V) 5 V, 15 V Min AD797A Typ 25 50 0.2 0.25 0.5 100 120 20 6 15 5 20,000 Max 80 125/180 1.0 1.5 3.0 400 600/700 Min AD797B Typ 10 30 0.2 0.25 0.25 80 120 2 20 2 10 2 15 2 7 14,000 20,000 Max 40 60 0.6 0.9 2.0 200 300 Unit V V V/C A A nA nA V/V V/V V/V V/V V/V
DYNAMIC PERFORMANCE Gain Bandwidth Product -3 dB Bandwidth Full Power Bandwidth1 Slew Rate Settling Time to 0.0015% COMMON-MODE REJECTION POWER SUPPLY REJECTION INPUT VOLTAGE NOISE
15 V 15 V 15 V 15 V 15 V 15 V 5 V, 15 V 12.5 114 110 114 110
110 450 8 280 20 800 130 120 130 120 50 1.7 0.9 1.0 2.0 12 3 13 13 3 80 50 -98 -120
110 450 8 280 12.5 1200 120 114 120 130 20 800 130 120 114 120 50 1.7 0.9 1.0 2.0 11 2.5 13 13 3 80 50 -98 -120
MHz MHz MHz kHz V/s ns dB dB dB dB nV p-p nV/Hz nV/Hz V rms pA/Hz V V V V V mA mA dB dB
1200
INPUT CURRENT NOISE INPUT COMMON-MODE VOLTAGE RANGE OUTPUT VOLTAGE SWING
RLOAD = 2 k RLOAD = 600 RLOAD = 600
Short-Circuit Current Output Current 3 TOTAL HARMONIC DISTORTION
RLOAD = 1 k, CN = 50 pF, f = 250 kHz, 3 V rms RLOAD = 1 k, f = 20 kHz, 3 V rms
15 V 15 V 15 V 15 V 15 V 15 V 5 V 15 V 15 V 5 V 5 V, 15 V 5 V, 15 V 15 V 15 V
1.2 1.3
2.5 1.2 1.2 12 3
11 2.5 12 11 2.5 30
12 11 2.5 30 -90 -110
-90 -110
INPUT CHARACTERISTICS Input Resistance Differential Common Mode Input Capacitance Differential 4 Common Mode
7.5 100 20 5
7.5 100 20 5
k M pF pF
Rev. F | Page 3 of 20
AD797
Parameter OUTPUT RESISTANCE POWER SUPPLY Operating Range Quiescent Current
1 2
Conditions AV = 1, f = 1 kHz
Supply Voltage (V)
Min
AD797A Typ Max 3 18 10.5
Min
AD797B Typ
Max 3 18 10.5
Unit m V mA
5 5 V, 15 V 8.2
5 8.2
Full power bandwidth = slew rate/2 VPEAK. Specified using external decompensation capacitor. 3 Output current for |VS - VOUT| > 4 V, AOL > 200 k. 4 Differential input capacitance consists of 1.5 pF package capacitance and 18.5 pF from the input differential pair.
Rev. F | Page 4 of 20
AD797 ABSOLUTE MAXIMUM RATINGS
Table 2.
Parameter Supply Voltage Internal Power Dissipation @ 25C1 PDIP SOIC Input Voltage Differential Input Voltage2 Output Short-Circuit Duration Ratings 18 V 1.3 W - (TA - 25C)/JA 0.9 W (TA - 25C)/JA VS 0.7 V Indefinite within maximum internal power dissipation -65C to +125C -40C to +85C 300C
Stresses above those listed under Absolute Maximum Ratings may cause permanent damage to the device. This is a stress rating only; functional operation of the device at these or any other conditions above those indicated in the operational section of this specification is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability.
ESD CAUTION
Storage Temperature Range (N, R Suffix) Operating Temperature Range Lead Temperature Range (Soldering 60 sec)
1 2
JA = 95C/W for the 8-lead PDIP; 155C/W for the 8-lead SOIC. The AD797 inputs are protected by back-to-back diodes. To achieve low noise, internal current-limiting resistors are not incorporated into the design of this amplifier. If the differential input voltage exceeds 0.7 V, the input current should be limited to less than 25 mA by series protection resistors. Note, however, that this degrades the low noise performance of the device.
Rev. F | Page 5 of 20
AD797 TYPICAL PERFORMANCE CHARACTERISTICS
20
INPUT COMMON-MODE RANGE (V)
15
10
5
VERTICAL SCALE (0.01V/DIV)
0
5
10 SUPPLY VOLTAGE (V)
15
20
00846-004
0
HORIZONTAL SCALE (5sec/DIV)
Figure 4. Input Common-Mode Voltage Range vs. Supply Voltage
Figure 7. 0.1 Hz to 10 Hz Noise
20
0
OUTPUT VOLTAGE SWING (V)
INPUT BIAS CURRENT (A)
15
-0.5
10
-VOUT +VOUT
-1.0
5
-1.5
00846-005
0
5
10 SUPPLY VOLTAGE (V)
15
20
-40
-20
0
20
40
60
80
100
120
140
TEMPERATURE (C)
Figure 5. Output Voltage Swing vs. Supply Voltage
Figure 8. Input Bias Current vs. Temperature
30 VS = 15V
140
OUTPUT VOLTAGE SWING (V p-p)
SHORT-CIRCUIT CURRENT (mA)
120
20
100 SINK CURRENT 80 SOURCE CURRENT
10 VS = 5
60
TEMPERATURE (C)
Figure 6. Output Voltage Swing vs. Load Resistance
Figure 9. Short-Circuit Current vs. Temperature
Rev. F | Page 6 of 20
00846-009
100 1k LOAD RESISTANCE ()
10k
00846-006
0 10
40 -60
-40
-20
0
20
40
60
80
100
120
140
00846-008
0
-2.0 -60
00846-007
AD797
11
140
200
QUIESCENT SUPPLY CURRENT (mA)
POWER SUPPLY REJECTION (dB)
10
+125C
100
PSR -SUPPLY
PSR +SUPPLY 150
9
8
+25C
80 CMR 60
125
100
7
-55C
6
40
75
1
10
100
1k
10k
100k
SUPPLY VOLTAGE (V)
FREQUENCY (Hz)
Figure 10. Quiescent Supply Current vs. Supply Voltage
Figure 13. Power Supply and Common-Mode Rejection vs. Frequency
12
-60
f = 1kHz RL = 600 G = +10
RL = 600 G = +10 f = 10kHz NOISE BW = 100kHz
9
OUTPUT VOLTAGE (V rms)
6
THD + NOISE (dB)
-80
VS = 5V -100
VS = 15V
3
0.1 OUTPUT LEVEL (V)
1
10
SUPPLY VOLTAGE (V)
Figure 11. Output Voltage vs. Supply Voltage for 0.01% Distortion
Figure 14. Total Harmonic Distortion (THD) + Noise vs. Output Level
1.0
30
15V SUPPLIES
0.8
SETTLING TIME (s)
0.0015%
OUTPUT VOLTAGE (V p-p)
RL = 600
20
0.6
0.01%
0.4
10
0.2
5V SUPPLIES
00846-012
0 0 2 4 6 8 10 STEP SIZE (V)
100k
1M
10M
FREQUENCY (Hz)
Figure 12. Settling Time vs. Step Size ()
Figure 15. Large-Signal Frequency Response
Rev. F | Page 7 of 20
00846-015
0 10k
00846-014
0
5
10
15
20
00846-011
0
-120 0.01
00846-013
0
5
10
15
20
00846-010
20
50 1M
COMMON MODE REJECTION (dB)
120
175
AD797
5
35
120
GAIN/BANDWIDTH PRODUCT (MHz (G = 1000))
00846-021 00846-020
00846-019
INPUT VOLTAGE NOISE (nV/Hz)
4
GAIN/BANDWIDTH PRODUCT 30
SLEW RATE (V/s)
110 SLEW RATE RISING EDGE 100
3
25
2
SLEW RATE FALLING EDGE 20 90
1
00846-016
0 10 100 1k 10k 100k 1M 10M FREQUENCY (Hz)
15 -60
-40
-20
0
20
40
60
80
100
120
80 140
TEMPERATURE (C)
Figure 16. Input Voltage Noise Spectral Density
Figure 19. Slew Rate and Gain/Bandwidth Product vs. Temperature
120
100
160
PHASE MARGIN
100
OPEN-LOOP GAIN (dB)
80
60
OPEN-LOOP GAIN (dB)
WITH RS*
PHASE MARGIN (Degrees)
WITHOUT RS*
80
140
60
40
GAIN
40 20
120
*RS = 100
20
WITHOUT RS* WITH RS*
0
*SEE FIGURE 25.
0 100 1k 10k
00846-017
100k 1M FREQUENCY (Hz)
10M
100M
100
100
1k LOAD RESISTANCE ()
10k
Figure 17. Open-Loop Gain and Phase Margin vs. Frequency
Figure 20. Open-Loop Gain vs. Load Resistance
300
100
INPUT OFFSET CURRENT (nA)
150
OVERCOMPENSATED
MAGNITUDE OF OUTPUT IMPEDANCE ()
10
0
1 WITHOUT CN*
-150 UNDER COMPENSATED
0.1
*SEE FIGURE 32. WITH CN*
TEMPERATURE (C)
00846-018
-300 -60
-40
-20
0
20
40
60
80
100
120
140
0.01
10
100
1k
10k
100k
1M
FREQUENCY (Hz)
Figure 18. Input Offset Current vs. Temperature
Figure 21. Magnitude of Output Impedance vs. Frequency
Rev. F | Page 8 of 20
AD797
20pF 1k +VS 1k
2
100 +VS
**
*
7
VIN
2
7
AD797
3 4
6
VOUT
VIN
RS*
AD797
3 4
6
VOUT 600
**
* -VS
*SEE FIGURE 35.
00846-022
-VS
00846-025
*VALUE OF SOURCE RESISTANCE (SEE THE NOISE AND SOURCE IMPEDANCE CONSIDERATIONS SECTION). **SEE FIGURE 35.
Figure 22. Inverter Connection
Figure 25. Follower Connection
1s
100 90
5V
100 90
1s
10
10
5V
Figure 23. Inverter Large-Signal Pulse Response
00846-023
Figure 26. Follower Large-Signal Pulse Response
50mV
100 90
100ns
100 90
50mV
100ns
10
00846-024
10
00846-027
0%
0%
Figure 24. Inverter Small-Signal Pulse Response
Figure 27. Follower Small-Signal Pulse Response
Rev. F | Page 9 of 20
00846-026
0%
0%
AD797
50mV
100 90
500ns
100 90
50mV
500ns
10
00846-028
10
00846-029
0%
0%
Figure 28. 16-Bit Settling Time Positive Input Pulse
Figure 29. 16-Bit Settling Time Negative Input Pulse
Rev. F | Page 10 of 20
AD797 THEORY OF OPERATION
The architecture of the AD797 was developed to overcome inherent limitations in previous amplifier designs. Previous precision amplifiers used three stages to ensure high open-loop gain (see Figure 30) at the expense of additional frequency compensation components. Slew rate and settling performance are usually compromised, and dynamic performance is not adequate beyond audio frequencies. As can be seen in Figure 30, the first stage gain is rolled off at high frequencies by the compensation network. Second stage noise and distortion then appears at the input and degrade performance. The AD797, on the other hand, uses a single ultrahigh gain stage to achieve dc as well as dynamic precision. As shown in the simplified schematic (Figure 31), Node A, Node B, and Node C track the input voltage, forcing the operating points of all pairs of devices in the signal path to match. By exploiting the inherent matching of devices fabricated on the same IC chip, high open-loop gain, CMRR, PSRR, and low VOS are guaranteed by pairwise device matching (that is, NPN to NPN and PNP to PNP), not by an absolute parameter such as beta and the early voltage.
gm R1 C1 BUFFER RL VOUT
benefit of making the low noise of the AD797 (<0.9 nV/Hz) extend to beyond 1 MHz. This means new levels of performance for sampled data and imaging systems. All of this performance as well as load drive in excess of 30 mA are made possible by the Analog Devices, Inc., advanced complementary bipolar (CB) process. Another unique feature of this circuit is that the addition of a single capacitor, CN (see Figure 31), enables cancellation of distortion due to the output stage. This can best be explained by referring to a simplified representation of the AD797 using idealized blocks for the different circuit elements (Figure 32). A single equation yields the open-loop transfer function of this amplifier; solving it at Node B yields
VOUT V IN
=
gm C CN j - C N j - C j A A
where: gm is the transconductance of Q1 and Q2. A is the gain of the output stage (~1). VOUT is voltage at the output. VIN is differential input voltage. When CN is equal to CC, the ideal single-pole op amp response is attained:
GAIN = gm x R1 x 5 x 106 a. C2
VOUT V IN
BUFFER RL
00846-030
=
gm jC
gm R1 C1 R2
A2
A3
VOUT
GAIN = gm x R1 x A2 x A3 b.
Figure 30. Model of AD797 vs. That of a Typical Three-Stage Amplifier
VCC R2 R3 CN Q4 Q3 A +IN Q1 Q2
-IN
In Figure 32, the terms of Node A, which include the properties of the output stage, such as output impedance and distortion, cancel by simple subtraction. Therefore, the distortion cancellation does not affect the stability or frequency response of the amplifier. With only 500 A of output stage bias, the AD797 delivers a 1 kHz sine wave into 60 at 7 V rms with only 1 ppm of distortion.
I1 I2 CN
R1 Q7 B
I5 Q10 Q9
A
VOUT
B A VOUT
Q5
Q6
CC
Q12
Q8 Q11 I6
+IN
-IN
Q1
Q2
CURRENT MIRROR
CC
1
00846-031
I1
I7
I4 VSS
I3
C
I4
Figure 31. AD797 Simplified Schematic
Figure 32. AD797 Block Diagram
This matching benefits not just dc precision, but, because it holds up dynamically, both distortion and settling time are also reduced. This single stage has a voltage gain of >5 x 106 and VOS < 80 V, while at the same time providing a THD + noise of less than -120 dB and true 16-bit settling in less than 800 ns. The elimination of second-stage noise effects has the additional
NOISE AND SOURCE IMPEDANCE CONSIDERATIONS
The AD797 ultralow voltage noise of 0.9 nV/Hz is achieved with special input transistors running at nearly 1 mA of collector current. Therefore, it is important to consider the total input-referred noise (eNtotal), which includes contributions
Rev. F | Page 11 of 20
00846-032
C
AD797
from voltage noise (eN), current noise (iN), and resistor noise (4 kTRS). The plot in Figure 7 uses a slightly different technique: an FFT-based instrument (Figure 34) is used to generate a 10 Hz "brickwall" filter. A low frequency pole at 0.1 Hz is generated with an external ac coupling capacitor, which is also the instrument being dc coupled. Several precautions are necessary to attain optimum low frequency noise performance: * Care must be used to account for the effects of RS. Even a 10 resistor has 0.4 nV/Hz of noise (an error of 9% when root sum squared with 0.9 nV/Hz). The test setup must be fully warmed up to prevent eOS drift from erroneously contributing to input noise. Circuitry must be shielded from air currents. Heat flow out of the package through its leads creates the opportunity for a thermoelectric potential at every junction of different metals. Selective heating and cooling of these by random air currents appears as 1/f noise and obscures the true device noise. The results must be interpreted using valid statistical techniques.
100k +VS *
1
2 7
e N total = [e N 2 + 4 kTR S + (i N x R S ) 2 ]1 / 2
where RS is the total input source resistance.
(1)
This equation is plotted for the AD797 in Figure 33. Because optimum dc performance is obtained with matched source resistances, this case is considered even though it is clear from Equation 1 that eliminating the balancing source resistance lowers the total noise by reducing the total RS by a factor of 2. At very low source resistance (RS < 50 ), the voltage noise of the amplifier dominates. As source resistance increases, the Johnson noise of RS dominates until a higher resistance of RS > 2 k is achieved; the current noise component is larger than the resistor noise.
100
* *
TOTAL NOISE 10
NOISE (nV/Hz)
*
1
RESISTOR NOISE ONLY
1.5F
6
AD797
3
00846-033
VOUT
0.1 10 100 1000 10000
SOURCE RESISTANCE ()
4
HP 3465 DYNAMIC SIGNAL ANALYZER (10Hz)
* -VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
00846-034
Figure 33. Noise vs. Source Resistance
The AD797 is the optimum choice for low noise performance if the source resistance is kept <1 k. At higher values of source resistance, optimum performance with respect to only noise is obtained with other amplifiers from Analog Devices (Table 3). Table 3. Recommended Amplifiers for Different Source Impedances
RS (k) 0 to <1 1 to <10 10 to <100 >100 Recommended Amplifier AD797 AD743/AD745, OP27/OP37, OP07 AD743/AD745, OP07 AD548, AD549, AD711, AD743/AD745
Figure 34. Test Setup for Measuring 0.1 Hz to 10 Hz Noise
WIDEBAND NOISE
Due to its single-stage design, the noise of the AD797 is flat over frequencies from less than 10 Hz to beyond 1 MHz. This is not true of most dc precision amplifiers, where second-stage noise contributes to input-referred noise beyond the audio frequency range. The AD797 offers new levels of performance in wideband imaging applications. In sampled data systems, where aliasing of out-of-band noise into the signal band is a problem, the AD797 outperforms all previously available IC op amps.
BYPASSING CONSIDERATIONS
Taking full advantage of the very wide bandwidth and dynamic range capabilities of the AD797 requires some precautions. First, multiple bypassing is recommended in any precision application. A 1.0 F to 4.7 F tantalum in parallel with 0.1 F ceramic bypass capacitors are sufficient in most applications. When driving heavy loads, a larger demand is placed on the supply bypassing. In this case, selective use of larger values of tantalum capacitors and damping of their lead inductance with small-value (1.1 to 4.7 ) carbon resistors can achieve an improvement. Figure 35 summarizes power supply bypassing recommendations.
LOW FREQUENCY NOISE
Analog Devices specifies low frequency noise as a peak-to-peak quantity in a 0.1 Hz to 10 Hz bandwidth. Several techniques can be used to make this measurement. The usual technique involves amplifying, filtering, and measuring the amplifier noise for a predetermined test time. The noise bandwidth of the filter is corrected for, and the test time is carefully controlled because the measurement time acts as an additional low frequency roll-off.
Rev. F | Page 12 of 20
AD797
VS VS
OR
0.1F 4.7F
0.1F
4.7F TO 22.0F
1.1 TO 4.7 KELVIN RETURN USE SHORT LEAD LENGTHS (<5mm) USE SHORT LEAD LENGTHS (<5mm) KELVIN RETURN
Low noise preamplification is usually performed in the noninverting mode (Figure 38). For lowest noise, the equivalent resistance of the feedback network should be as low as possible. The 30 mA minimum drive current of the AD797 makes it easier to achieve this. The feedback resistors can be made as low as possible, with consideration to load drive and power consumption.
00846-035
LOAD CURRENT
LOAD CURRENT
CL
Figure 35. Recommended Power Supply Bypassing
R2 +VS * R1
2 7
THE NONINVERTING CONFIGURATION
Ultralow noise requires very low values of the internal parasitic resistance (rBB) for the input transistors (6 ). This implies very little damping of input and output reactive interactions. With the AD797, additional input series damping is required for stability with direct output to input feedback. A 100 resistor (R1) in the inverting input (Figure 36) is sufficient; the 100 balancing resistor (R2) is recommended but is not required for stability. The noise penalty is minimal (eNtotal 2.1 nV/Hz), which is usually insignificant.
R1 100 +VS *
2 7
AD797
VIN
3 4
6
VOUT
RL
* -VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
00846-038
Figure 38. Low Noise Preamplifier
VIN
R2 100
AD797
3 4
6
RL 600 *
VOUT
Table 4 provides some representative values for the AD797 when used as a low noise follower. Operation on 5 V supplies allows the use of a 100 or less feedback network (R1 + R2). Because the AD797 shows no unusual behavior when operating near its maximum rated current, it is suitable for driving the AD600/AD602 (see Figure 50) while preserving low noise performance. Optimum flatness and stability at noise gains >1 sometimes require a small capacitor (CL) connected across the feedback resistor (R1 of Figure 38). Table 4 includes recommended values of CL for several gains. In general, when R2 is greater than 100 and CL is greater than 33 pF, a 100 resistor should be placed in series with CL. Source resistance matching is assumed, and the AD797 should not be operated with unbalanced source resistance >200 k/G. Table 4. Values for Follower with Gain Circuit
Gain 2 2 10 20 >35 R1 1 k 300 33.2 16.5 10 R2 1 k 300 300 316 (G - 1) x 10 CL 20 pF 10 pF 5 pF Noise (Excluding RS) 3.0 nV/Hz 1.8 nV/Hz 1.2 nV/Hz 1.0 nV/Hz 0.98 nV/Hz
-VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
Figure 36. Voltage Follower Connection
Best response flatness is obtained with the addition of a small capacitor (CL < 33 pF) in parallel with the 100 resistor (Figure 37). The input source resistance and capacitance also affect the response slightly, and experimentation may be necessary for best results.
CL
100 +VS *
2 7
VIN
RS CS
AD797
3 4
6
VOUT
600
00846-036
* -VS
00846-037
*USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
Figure 37. Alternative Voltage Follower Connection
The I-to-V converter is a special case of the follower configuration. When the AD797 is used in an I-to-V converter, for example as a DAC buffer, the circuit shown in Figure 39 should be used. The value of CL depends on the DAC, and if CL is greater than 33 pF, a 100 series resistor is required. A bypassed balancing resistor (RS and CS) can be included to minimize dc errors.
Rev. F | Page 13 of 20
AD797
20pF TO 120pF R1 +VS IIN
2 7
100
DRIVING CAPACITIVE LOADS
The capacitive load driving capabilities of the AD797 are displayed in Figure 41. At gains greater than 10, usually no special precautions are necessary. If more drive is desirable, however, the circuit shown in Figure 42 should be used. For example, this circuit allows a 5000 pF load to be driven cleanly at a noise gain 2.
100nF
*
AD797
3 4
6
VOUT
600
CS
RS -VS
*
CAPACITIVE LOAD DRIVE CAPABILITY
00846-039
10nF
*USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
Figure 39. I-to-V Converter Connection
THE INVERTING CONFIGURATION
The inverting configuration (see Figure 40) presents a low input impedance, R1, to the source. For this reason, the goals of both low noise and input buffering are at odds with one another. Nonetheless, the excellent dynamics of the AD797 makes it the preferred choice in many inverting applications, and with careful selection of feedback resistors, the noise penalties are minimal. Some examples are presented in Table 5 and Figure 40.
CL R2
1nF
100pF
10pF
1
10
100
1k
CLOSED-LOOP GAIN
Figure 41. Capacitive Load Drive Capability vs. Closed-Loop Gain
20pF 1k
+VS * R1
2
200pF +VS
6
100
VIN
7
AD797
3 4
VOUT
* 1k
2 7
RL *
00846-040
VIN
3
RS -VS
AD797
4
33
6
VOUT
C1
* -VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
00846-042
*USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
Figure 40. Inverting Amplifier Connection
Table 5. Values for Inverting Circuit
Gain -1 -1 -10 R1 1 k 300 150 R2 1 k 300 1500 CL 20 pF 10 pF 5 pF Noise (Excluding RS) 3.0 nV/Hz 1.8 nV/Hz 1.8 nV/Hz
Figure 42. Recommended Circuit for Driving a High Capacitance Load
SETTLING TIME
The AD797 is unique among ultralow noise amplifiers in that it settles to 16 bits (<150 V) in less than 800 ns. Measuring this performance presents a challenge. A special test circuit (see Figure 43) was developed for this purpose. The input signal was obtained from a resonant reed switch pulse generator, available from Tektronix as calibration Fixture No. 067-0608-00. When open, the switch is simply 50 to ground and settling is purely a passive pulse decay and inherently flat. The low repetition rate signal was captured on a digital oscilloscope after being amplified and clamped twice. The selection of plug-in for the oscilloscope was made for minimum overload recovery.
Rev. F | Page 14 of 20
00846-041
1pF
AD797
TO TEKTRONIX 7A26 OSCILLOSCOPE 1M PREAMP INPUT SECTION 226 4.26k (VIA LESS THAN 1FT 50 COAXIAL CABLE)
2
20pF
The benefits of adding C1 are evident for closed-loop gains of 100. A maximum value of 33 pF at gains of 1000 is recommended. At a gain of 1000, the bandwidth is 450 kHz. Table 6 and Figure 45 summarize the performance of the AD797 with distortion cancellation and decompensation.
R1
- A2
AD829
3
6 7
250
VERROR x 5
2x HP2835
50pF R2
2 8
+
4
2x HP2835
0.47F 0.47F +VS -VS 1k 1k 100
VIN
AD797
3
6
a.
TEKTRONIX CALIBRATION FIXTURE
1k
R1
VIN
1k
2
20pF - +
4
A1
6 7
C2 R2 C1
2 8
AD797
3
51pF
AD797
1F 0.1F +VS -VS
00846-043
6
VOUT
1F
0.1F
VIN
3
NOTES USE CIRCUIT BOARD WITH GROUND PLANE.
b.
C1, SEE TABLE C2 = 50pF - C1
Figure 43. Settling Time Test Circuit
Figure 44. Recommended Connections for Distortion Cancellation and Bandwidth Enhancement
DISTORTION REDUCTION
The AD797 has distortion performance (THD < -120 dB, @ 20 kHz, 3 V rms, RL = 600 ) unequaled by most voltage feedback amplifiers. At higher gains and higher frequencies, THD increases due to a reduction in loop gain. However, in contrast to most conventional voltage feedback amplifiers, the AD797 provides two effective means of reducing distortion as gain and frequency are increased: cancellation of the output stage's distortion and gain bandwidth enhancement by decompensation. By applying these techniques, gain bandwidth can be increased to 450 MHz at G = 1000, and distortion can be held to -100 dB at 20 kHz for G = 100. The unique design of the AD797 provides cancellation of the output stage's distortion. To achieve this, a capacitance equal to the effective compensation capacitance, usually 50 pF, is connected between Pin 8 and the output (see C2 in Figure 44). Use of this feature improves distortion performance when the closed-loop gain is more than 10 or when frequencies of interest are greater than 30 kHz. Bandwidth enhancement via decompensation is achieved by connecting a capacitor from Pin 8 to ground (see C1 in Figure 44). Adding C1 results in subtracting from the value of the internal compensation capacitance (50 pF), yielding a smaller effective compensation capacitance and therefore a larger bandwidth.
Table 6. Recommended External Compensation for Distortion Cancellation and Bandwidth Enhancement
A/B R1 R2 () () 909 100 1k 10 10 k 10 A C1 (pF) 0 0 0 C2 (pF) 50 50 50 3 dB BW 6 MHz 1 MHz 110 kHz C1 (pF) 0 15 33 B C2 (pF) 50 33 15 3 dB BW 6 MHz 1.5 MHz 450 kHz
0.01 G = +1000 RL = 600 -90 NOISE LIMIT, G = +1000 G = +1000 RL = 10k G = +100 RL = 600 NOISE LIMIT, G = +100 -110 0.003
Gain 10 100 1000
-80
00846-044
THD (dB)
-100
0.001
0.0003
-120
G = +10 RL = 600
0.0001
00846-045
100
300
1k
3k
10k
30k
100k
300k
FREQUENCY (Hz)
Figure 45. Total Harmonic Distortion (THD) vs. Frequency @ 3 V rms for Figure 44b
Differential Line Receiver
The differential receiver circuit of Figure 46 is useful for many applications, from audio to MRI imaging. The circuit allows
Rev. F | Page 15 of 20
THD (%)
AD797
extraction of a low level signal in the presence of commonmode noise. As shown in Figure 47, the AD797 provides this function with only 9 nV/Hz noise at the output. Figure 48 shows the AD797 20-bit THD performance over the audio band and the 16-bit accuracy to 250 kHz.
20pF 1k +VS DIFFERENTIAL INPUT
7 2 8
A General-Purpose ATE/Instrumentation I/O Driver
The ultralow noise and distortion of the AD797 can be combined with the wide bandwidth, slew rate, and load drive of a current feedback amplifier to yield a very wide dynamic range general-purpose driver. The circuit shown in Figure 49 combines the AD797 with the AD811 in just such an application. Using the component values shown, this circuit is capable of better than -90 dB THD with a 5 V, 500 kHz output signal. The circuit is, therefore, suitable for driving a high resolution ADC as an output driver in automatic test equipment (ATE) systems. Using a 100 kHz sine wave, the circuit drives a 600 load to a level of 7 V rms with less than -109 dB THD and a 10 k load at less than -117 dB THD.
22pF R2 2k +VS * +VS
1k ** 50pF*
AD797
3 4
6
VOUT
** 1k 1k
-VS
20pF * OPTIONAL ** USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
00846-046
2
7
*
AD797
VIN 1k
3 4
6
3
7
Figure 46. Differential Line Receiver
16
AD811
*
2 4
6
VOUT
-VS
*
OUTPUT VOLTAGE NOISE (nV/Hz)
14
649 649
-VS
12
*USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
10
Figure 49. A General-Purpose ATE/Instrumentation I/O Driver
8
Ultrasound/Sonar Imaging Preamp
The AD600 variable gain amplifier provides the time-controlled gain (TCG) function necessary for very wide dynamic range sonar and low frequency ultrasound applications. Under some circumstances, it is necessary to buffer the input of the AD600 to preserve its low noise performance. To optimize dynamic range, this buffer should have a maximum of 6 dB of gain. The combination of low noise and low gain is difficult to achieve. The input buffer circuit shown in Figure 50 provides 1 nV/Hz noise performance at a gain of 2 (dc to 1 MHz) by using 26.1 resistors in its feedback path. Distortion is only -50 dBc at 1 MHz for a 2 V p-p output level and drops rapidly to better than -70 dBc at an output level of 200 mV p-p.
6
FREQUENCY (Hz)
00846-047
10
100
1k
10k
100k
1M
10M
Figure 47. Output Voltage Noise Spectral Density for Differential Line Receiver
-90 0.003
-100
WITHOUT OPTIONAL 50pF CN
0.001
THD (dB)
-110
MEASUREMENT LIMIT
0.0003
-120 WITH OPTIONAL 50pF CN 300 1k 3k 10k 30k 100k
0.0001
300k
FREQUENCY (Hz)
Figure 48. Total Harmonic Distortion (THD) vs. Frequency for Differential Line Receiver
Rev. F | Page 16 of 20
00846-048
-130 100
THD (%)
00846-049
AD797
26.1 +VS * 26.1
2 7
Professional Audio Signal Processing--DAC Buffers
*
AD797
VIN
3 4
6
AD600
* VS = 6Vdc
VOUT
* -VS
*USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
Figure 50. An Ultrasound Preamplifier Circuit
Amorphous (Photodiode) Detector
Large area photodiodes (CS 500 pF) and certain image detectors (amorphous Si) have optimum performance when used in conjunction with amplifiers with very low voltage (rather than very low current noise). Figure 51 shows the AD797 used with an amorphous Si (CS = 1000 pF) detector. The response is adjusted for flatness using capacitor CL, and the noise is dominated by voltage noise amplified by the ac noise gain. The AD797's excellent input noise performance gives 27 V rms total noise in a 1 MHz bandwidth, as shown by Figure 52.
100 CL 50pF
The low noise and low distortion of the AD797 make it an ideal choice for professional audio signal processing. An ideal I-to-V converter for a current output DAC would simply be a resistor to ground, were it not for the fact that most DACs do not operate linearly with voltage on their output. Standard practice is to operate an op amp as an I-to-V converter, creating a virtual ground at its inverting input. Normally, clock energy and current steps must be absorbed by the op amp output stage. However, in the configuration shown in Figure 53, Capacitor CF shunts high frequency energy to ground while correctly reproducing the desired output with extremely low THD and IMD.
CF 82pF 100
00846-050
3k +VS *
AD1862 DAC
2
7
C1 2000pF
3
AD797
4
6
VOUT
* -VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
00846-053
10k +VS *
2 7
Figure 53. A Professional Audio DAC Buffer
+VS
-IN
6
2
7
IS
CS 1000pF
3
AD797
4
VOUT
+IN
00846-051
AD797
3 1 4 5
6
VOUT
* -VS *USE THE POWER SUPPLY BYPASSING SHOWN IN FIGURE 35.
20k
00846-054
VOS ADJUST -VS
Figure 51. Amorphous Detector Preamp
VOLTAGE NOISE (mV rms (0.1Hz FREQUENCY))
-30
100
Figure 54. Offset Null Configuration
-40
VOUT (dB Re 1V/A)
VOUT
80 NOISE 60
-50
-60
40
-70
20
100
1k
10k
100k 1M FREQUENCY (Hz)
10M
Figure 52. Total Integrated Voltage Noise and VOUT of Amorphous Detector Preamp
Rev. F | Page 17 of 20
00846-052
-80
0 100M
AD797 OUTLINE DIMENSIONS
0.400 (10.16) 0.365 (9.27) 0.355 (9.02)
8 1 5
4
0.280 (7.11) 0.250 (6.35) 0.240 (6.10)
0.100 (2.54) BSC 0.210 (5.33) MAX 0.150 (3.81) 0.130 (3.30) 0.115 (2.92) 0.022 (0.56) 0.018 (0.46) 0.014 (0.36) 0.070 (1.78) 0.060 (1.52) 0.045 (1.14)
0.325 (8.26) 0.310 (7.87) 0.300 (7.62) 0.060 (1.52) MAX 0.195 (4.95) 0.130 (3.30) 0.115 (2.92)
0.015 (0.38) MIN SEATING PLANE 0.005 (0.13) MIN
0.015 (0.38) GAUGE PLANE 0.430 (10.92) MAX
0.014 (0.36) 0.010 (0.25) 0.008 (0.20)
COMPLIANT TO JEDEC STANDARDS MS-001 CONTROLLING DIMENSIONS ARE IN INCHES; MILLIMETER DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF INCH EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN. CORNER LEADS MAY BE CONFIGURED AS WHOLE OR HALF LEADS.
Figure 55. 8-Lead Plastic Dual In-Line Package [PDIP] Narrow Body (N-8) Dimensions shown in inches and (millimeters)
5.00 (0.1968) 4.80 (0.1890)
4.00 (0.1574) 3.80 (0.1497)
8 1
5 4
6.20 (0.2441) 5.80 (0.2284)
1.27 (0.0500) BSC 0.25 (0.0098) 0.10 (0.0040) COPLANARITY 0.10 SEATING PLANE
1.75 (0.0688) 1.35 (0.0532)
0.50 (0.0196) 0.25 (0.0099) 8 0 0.25 (0.0098) 0.17 (0.0067) 1.27 (0.0500) 0.40 (0.0157)
45
0.51 (0.0201) 0.31 (0.0122)
COMPLIANT TO JEDEC STANDARDS MS-012-A A CONTROLLING DIMENSIONS ARE IN MILLIMETERS; INCH DIMENSIONS (IN PARENTHESES) ARE ROUNDED-OFF MILLIMETER EQUIVALENTS FOR REFERENCE ONLY AND ARE NOT APPROPRIATE FOR USE IN DESIGN.
Figure 56. 8-Lead Standard Small Outline Package [SOIC_N] Narrow Body (R-8) Dimensions shown in millimeters and (inches)
Rev. F | Page 18 of 20
012407-A
070606-A
AD797
ORDERING GUIDE
Model AD797AN AD797ANZ 1 AD797AR AD797AR-REEL AD797AR-REEL7 AD797ARZ1 AD797ARZ-REEL1 AD797ARZ-REEL71 AD797BR AD797BR-REEL AD797BR-REEL7 AD797BRZ1 AD797BRZ-REEL1 AD797BRZ-REEL71
1
Temperature Range -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C -40C to +85C
Package Description 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Plastic Dual In-Line Package [PDIP] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N] 8-Lead Standard Small Outline Package [SOIC_N]
Package Option N-8 N-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8 R-8
Z = RoHS Compliant Part.
Rev. F | Page 19 of 20
AD797 NOTES
(c)2008 Analog Devices, Inc. All rights reserved. Trademarks and registered trademarks are the property of their respective owners. D00846-0-1/08(F)
Rev. F | Page 20 of 20

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